This application is related to power conversion snubber circuits.
Switching power supplies gained popularity in the early 1970s, coinciding with the introduction of the bipolar power transistor. Since that time, many evolutionary changes have occurred to make the switching power supply meet the needs of many diverse applications. The boundaries to these application areas are determined primarily by the amount of stress the power switches (power transistors or MOSFETS) must endure and still provide reliable performance.
Background: DC--DC Converters and Single-Phase Fed AC-DC Converters
Power conversion, such as DC--DC and AC-DC was typically performed by hard-switched, pulse width modulated ("PWM") circuits such as the "boost" power converter shown in FIG. 1. It consists of an input ac source V1, an boost inductor L1, a boost switch S1, a boost diode D1, an output filter capacitor Co, and a load Ro. By varying the switching signal from the PWM controller, the duty cycle of S1 is varied and the output power is thus controlled. This is known as PWM control.
In the conventional boost power converter, when the switch S1 is open, current flows through the inductor L1 and the diode D1. The current flowing through inductor L1 and diode D1 will charge the capacitor C1. When the switch S1 is closed, current from the input voltage V1 will flow through L1 and S1 to the input voltage source. Assuming that there are no losses in the inductor L1 and negligible current drawn by the load, equating the volt-seconds across the inductor L1 to zero, and ignoring the turn-on voltage of D1, the maximum output voltage Vout can be determined from the following relationship: EQU Vout=V1/(1-D);
where D is the duty cycle of the switch.
Thus, assuming that a fixed-frequency switching signal is provided to the controllable switch S1, the higher the duty cycle of the switching signal, the higher the output voltage supplied across the output load Ro.
The above operation of the boost converter is discussed under the constraints that the power switch closely approximates that of ideal switches, i.e. that the transitions from open-to-closed and closed-to-open occur instantaneously. Unfortunately, this assumption is not particularly accurate, and consequently the finite turn-on time and finite turn-off time of the switching element causes switching losses of approximately P.apprxeq.I.sub.switch V.sub.switch.
A further problem is that, when the switch S1 turns off, the rate of rise of voltage dV/dt across S1 creates substantial unwanted radio-frequency interference ("RFI") and electromagnetic interference ("EMI") noise, which significantly limits the power conversion frequency and efficiency, particularly in the multi-kW power range.
In order to increase the power densities of the converters, the operating frequency needs to be as high as possible for any given power level. The biggest problems with conventional hard-switching PWM converters at multi-kilowatt power levels are attributable to diode stored charge, diode reverse recovery, and device switching losses. Therefore, conventional PWM converters operated at higher switching frequencies cause increased switching losses.
Background: Diode Reverse Recovery
Diode reverse recovery in the conventional hard switching boost power factor converter presents a significant limitation because it generates substantial EMI and limits the power conversion frequency and efficiency in multi-kilowatt power range. More specifically, in the conventional hard-switching boost converter of FIG. 1, when switch S1 is closed, the current through S1 increases to the level of the current through the inductor L1. At this point, the current through the diode D1 decreases until the diode D1 no longer conducts. At this time, any charge stored on the diode D1 is removed via switch S1. As the charge is being removed from the diode D1, the current through S1 continues to rise, often to a value of more than twice the inductor current level. The combination of high peak current (dI/dt) and high peak voltage (dV/dt) creates significant unwanted RFI/EMI noise, and considerably stresses the switch S1.
Background: "Soft-Switching" Reactive Snubber Circuits
In response to the problems associated with hard switching PWM power converters described above, designers have proposed the use of reactive snubber networks to "trap" energy that would normally be dissipated during switching transitions, and thereby permitting "soft-switching" of the controllable switch. Soft-switching occurs when little or no voltage appears across the switch and/or when little or no current is flowing through the switch, thereby reducing the switching stresses. Two types of such soft-switching converters are zero current switching ("ZCS") converters (in which turn-on and turn-off of switch occurs with no current in the controllable switch), and zero voltage switching ("ZVS") converters (in which turn-on and turn-off of the switch occurs with no voltage across it).
ZVS is accomplished by employing a purely capacitive snubber having an anti-parallel diode. The capacitor starts at an initial voltage, which places zero volts across the power switch at turn-off, and its voltage cannot be changed instantaneously. With ZVS circuits, turn-on occurs only when the anti-parallel diode is conducting, and turn-off losses decrease with increasing capacitance. Unfortunately, to cause conduction of the anti-parallel diode before turning-on the switch, additional circuitry is required to discharge the capacitor, or an external resistor is required to dissipate the energy stored in the capacitor during turn-on of switching cycle. Therefore, although the controllable switch can operate at elevated frequencies due to ZVS, the power losses are merely shifted from the switch to the dissipating resistor. Furthermore, if a snubber circuit is not properly designed, turn-on and turn-off of the switch itself can cause large current spikes.
On the other hand, ZCS is accomplished generally by employing a purely inductive snubber. The current through the inductor starts at an initial zero condition and lacks the ability to change instantaneously.
To solve the power dissipation problem, designers have developed "loss-less" soft-switching power converters in which the snubber networks are reset by means of inherent circuit operation. These loss-less soft-switching power converters "re-circulate" the energy stored by the reactive snubbers to accomplish loss-less operation. Unfortunately, this way of eliminating spikes requires extra power handling components which add size, weight, and cost to the power conversion system. Moreover, they often severely reduce overall system efficiency since the RMS input current is high.
Resonant Switching Converters
Other known loss-less power converters include resonant and quasi-resonant) switching converters. These converters incorporate reactive elements (capacitors and inductors) in conjunction with the switching device. In the purest sense of the word, resonance implies a continuous function whose waveform is a continuous sinusoidal signal. However, most so-called "resonant" converters are actually quasi-resonant converters--the resonant elements being operated for only one-half of a resonant sine wave at a time. The power switch in a quasi-resonant converter connects the input voltage source to the tank circuit, and is turned on and off in the same step fashion as a PWM switching power supply. The output voltage of these circuits is controlled by varying the operating frequency of the controllable switches. The conduction period (or "on" period) is dictated by the ringing frequency of the tank circuit. The power switch turns off after completion of one-half of a resonant period. A major advantage is that current during turn-on and turn-off is zero, therefore eliminating any switching losses. These circuits advantageously have low semiconductor switching losses and operate with sinusoidal waveforms. Unfortunately, resonant power converters exhibit increased component count, increased switching currents (peak and RMS), and require wide operating frequency variations to maintain a constant output voltage. Thus, resonant switching converters are relatively expensive, require complex switching control circuitry, and eliminate switching losses at the expense of conduction losses.
Modified Boost Power Converter
An example of a known ZVS resonant boost power converter is shown in FIG. 2. Further discussion regarding this circuit may be found in "Analysis and Design of a zero voltage transition power factor correction circuit," IEEE 1994 APEC, pp. 591-597, and is herein incorporated by reference. This is a modification of the hard-switching boost converter of FIG. 1. Specifically, a capacitor C2 is arranged across the switch S1. To achieve ZVS of S1, the auxiliary switch S2 is turned-on before turning-on S1. Then the current in the inductor L2 increases until it reaches the level of the current in L1. Simultaneously, the capacitor C2 and the inductor L2 create a resonance thereby reducing the voltage across the switch S1 to zero before S1 is turned-on. The diode D1 is turned off without the problem of a high reverse recovery current passing through the switch S1. The capacitor C2 minimizes the voltage across the switch S1 to a very low value during turn-off. Diode D4 blocks any current through diode D1 from entering the switching path of switch S2.
Unfortunately, the ZVS boost converter turn-off losses of the switch S2 are significant because the inductor L2 and S2 are carrying the load current before S2 is turned-off. Therefore, (1) the energy stored in the parasitic capacitance of the switch S2 dissipates in S2 during turn-on, (2) the switch S2 experiences substantial turnoff losses because it carries the full inductor current, (3) the inductor L2 must be designed to limit any reverse recovery current spike from the diode D1 during turn-on of S2, and (4) the tailing effect of IGBTs (Insulated Gate Bipolar Transistors) during turn-off causes difficulties when using IGBTs in power converters in a 1-5 kW power range.
Another example of a ZVS quasi-resonant boost power converter is shown in FIG. 3. During the turn-off of the switch S1, the inductor L2 and capacitor C2 cause a resonance, thereby making the voltage across the switch S1 nearly zero. Moreover, the switch S1 is closed when its anti-parallel diode D2 is conducting. Thus, the current is zero during turn-on of the switch S1. However, the current through S1 is sinusoidal. Thus, the peak and RMS currents are increased. Consequently, the quasi-resonant power converter eliminates switching losses at the expense of conduction losses. Furthermore, a wide range of operating switching frequencies is required to maintain a constant output voltage.
Summary of Conventional Prior-Art Circuits
Although these known loss-less boost power converters have better characteristics than the hard-switching converters, certain disadvantages still remain. Specifically, the known ZVS boost power converter second switch is not loss-less, as explained earlier. On the other hand, although a resonant switching power converter may eliminate switching losses, it increases conduction losses. Moreover, the output voltage in the resonant switching converters is controlled by frequency modulation. Therefore, this circuit requires additional circuitry to carry out such frequency modulation.
Accordingly, an improved switching power converter is needed for 1-5 kW output power levels. The improved active network should be capable of operating at higher switching frequencies to increase power density (watts per cubic inch). The improved snubber network should eliminate reverse recovery current spikes in the switching power converter. It should also limit peak device voltage and current stresses, limit peak capacitor voltages, limit RMS currents, have low sensitivity to second order effects, have low EMI, and be capable of using widely available control integrated circuits. Lastly, the improved active snubber network should be adaptable for use in various switching power converter circuits such as boost, buck, forward and flyback power converters.
Soft-Switched Built-In Active Snubber Circuit
This application discloses a zero-voltage soft-switched power converter using an active snubber circuit. The innovative soft-switching power converter has relatively low conduction losses compared to resonant topologies. This is accomplished by modifying conventional switching power converters and providing an improved active snubber network. The low losses are due to the capability of switching a pair of active-element networks simultaneously. Moreover, only one inductor is required with the pair of active networks.
This active snubber circuit improves efficiency, power density, and transient performance, reduces switching losses and EMI, and also operates at a fixed switching frequency. The proposed network also reduces and/or eliminates the large currents and reverse recovery current spikes normally seen in conventional switching power converters. This circuit may be used in various switching power topologies such as boost, flyback and forward power converters.